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Final Electrical Specifications
LTC1771 Low Quiescent Current High Efficiency Step-Down DC/DC Controller
FEATURES
s s s s s s s s s
DESCRIPTIO
February 2000
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Very Low Standby Current: 10A Available in Space-Saving 8-Lead MSOP Package High Output Currents Wide VIN Range: 2.8V to 20V Operation VOUT Range: 1.23V to 18V High Efficiency: Over 93% Possible 2% Output Accuracy Very Low Dropout Operation: 100% Duty Cycle Current Mode Operation for Excellent Line and Load Transient Response Defeatable Burst ModeTM Operation Short-Circuit Protected Optional Programmable Soft-Start Micropower Shutdown: IQ = 2A
The LTC(R)1771 is a high efficiency current mode stepdown DC/DC controller that draws as little as 10A DC supply current to regulate the output at no load while maintaining high efficiency for loads up to several amps. The LTC1771 drives an external P-channel power MOSFET using a current mode, constant off-time architecture. An external sense resistor is used to program the operating current level. Current mode control provides short-circuit protection, excellent transient response and controlled start-up behavior. Burst Mode operation enables the LTC1771 to maintain high efficiency down to extremely low currents. Shutdown mode further reduces the supply current to a mere 2A. For low noise applications, Burst Mode operation can be easily disabled with the MODE pin. Wide input supply range of 2.8V to 18V (20V maximum) and 100% duty cycle operation for low dropout make the LTC1771 ideal for a wide variety of battery-powered applications where maximizing battery life is important. The LTC1771's availability in both 8-lead MSOP and SO packages provides for a minimum area solution.
, LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation.
APPLICATIO S
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Cellular Telephones and Wireless Modems 1- to 4-Cell Lithium-Ion-Powered Applications Portable Instruments Battery-Powered Equipment Battery Chargers Scanners
TYPICAL APPLICATIO
VIN 4.5V TO 18V RSENSE 0.05 VIN RUN/SS CSS 0.01F RC 10k CC 22OpF SENSE PGATE M1 Si6447DQ 10F 25V CER
100 VIN = 5V 90
EFFICIENCY (%)
ITH LTC1771 VFB MODE GND VIN R2 1.64M 1%
L1 15H UPS5817
80 70 60 50
+
COUT 150F 6.3V
VOUT 3.3V 2A
R1 1M 1% 5pF
1771 F01
VOUT = 3.3V RSENSE = 0.05 40 10 0.1 1 100 1000 LOAD CURRENT (mA)
Figure 1. High Efficiency Step-Down Converter
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
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LTC1771 Efficiency
VIN = 10V VIN = 15V
10000
1771 F01b
1
LTC1771
ABSOLUTE AXI U RATI GS (Note 1)
Junction Temperature (Note 2) ............................ 125C Operating Temperature Range (Note 3) LTC1771E ......................................... - 40C to 85C LTC1771I ......................................... - 40C to 85C Storage Temperature Range ................. - 65C to 150C Lead Temperature (Soldering, 10 sec).................. 300C Input Supply Voltage (VIN)........................ - 0.3V to 20V Peak Driver Output Current < 10s (PGATE) ............. 1A RUN/SS Voltage ........................... - 0.3V to (VIN + 0.3V) MODE Voltage .......................................... - 0.3V to 20V ITH, VFB Voltage .......................................... - 0.3V to 5V SENSE Voltage (VIN > 12V)...(VIN - 12V) to (VIN + 0.3V) SENSE Voltage (VIN 12V) .......... - 0.3V to (VIN + 0.3V)
PACKAGE/ORDER I FOR ATIO
TOP VIEW RUN/SS ITH VFB GND 1 2 3 4 8 7 6 5 MODE SENSE VIN PGATE
ORDER PART NUMBER LTC1771EMS8 MS8 PART MARKING LTKD
RUN/SS 1 ITH 2 VFB 3 GND 4
MS8 PACKAGE 8-LEAD PLASTIC MSOP
TJMAX = 125C, JA = 150C/ W
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
The q denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25C. VIN = 10V, VRUN = open unless otherwise specified.
SYMBOL VFB IFB ISUPPLY VLINEREG VLOADREG IQ PARAMETER Feedback Voltage Feedback Current No-Load Supply Current Reference Voltage Line Regulation Output Voltage Load Regulation Input DC Supply Current Active Mode (PGATE = 0V) Sleep Mode (Note 6) Shutdown Short Circuit Maximum Current Sense Threshold Minimum Current Sense Threshold Sleep Current Sense Threshold Switch Off Time Mode Pin Threshold CONDITIONS (Note 5) (Note 5) VIN = 10V, ILOAD = 0 (Note 6) VIN = 5V to 15V (Note 5) ITH = 0.5V to 2V, Burst Disabled (Note 5) (Note 4) VIN = 2.8V to 18V VIN = 2.8V to 18V, VFB = 1.5V VIN = 2.8V to 18V, VRUN = 0V VIN = 2.8V to 18V, VFB = 0V VFB = VREF - 20mV VFB = VREF + 10mV, Burst Disabled ITH = 1V VFB at Regulated Value VFB = 0V VMODE Rising
q q q q q q
VSENSE(MAX) VSENSE(MIN) VSENSE(SLEEP) t OFF VMODE
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TOP VIEW 8 7 6 5 MODE SENSE VIN PGATE
ORDER PART NUMBER LTC1771ES8 LTC1771IS8 S8 PART MARKING 1771 1771I
S8 PACKAGE 8-LEAD PLASTIC SO
TJMAX = 125C, JA = 110C/ W
MIN 1.205
TYP 1.230 1 10 0.003 0.25 150 9 2 175
MAX 1.255 10 0.03 1 235 15 6 275 180
UNITS V nA A %/V % A A A A mV mV mV s s
110
140 - 25 50 3.5 70
0.5
1.3
2
V
LTC1771
ELECTRICAL CHARACTERISTICS
The q denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25C. VIN = 10V, VRUN = open unless otherwise specified.
SYMBOL VRUN/SS IRUN PGATE t r, tf PARAMETER RUN/SS Pin Threshold Source Current PGATE Transition Time (Note 7) Rise Time Fall Time CONDITIONS VRUN/SS Rising VRUN = 0V, VIN = 2.8V to 18V CLOAD = 2000pF CLOAD = 2000pF
q
MIN 0.5 0.3
TYP 1.0 1 80 90
MAX 2 3
UNITS V A ns ns
Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formulas: LTC1771S8: TJ = TA + (PD)(110C/W) LTC1771MS8: TJ = TA + (PD)(150C/W) Note 3: The LTC1771E is guaranteed to meet performance specifications from 0C to 70C. Specifications over the - 40C to 85C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LTC1771I is guaranteed and tested over the - 40C to 85C operating temperature range.
Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information. Note 5: The LTC1771 is tested in a feedback loop that servos VFB to the balance point for the error amplifier (VITH = 1.23V). Note 6: No-load supply current consists of sleep mode current (9A typical) plus a small switching component necessary to overcome Schottky diode leakage and feedback resistor current. Note 7: tr and tf measured at 10% to 90% levels.
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RUN/SS (Pin 1): The voltage level on this pin controls shutdown/run mode (ground = shutdown, open/high = run). Connecting an external capacitor to this pin provides soft-start. ITH (Pin 2): Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. Nominal voltage range for this pin is 0V to 3V. VFB (Pin 3): Feedback of Output Voltage for Comparison to Internal 1.23V Reference. An external resistive divider across the output is returned to this pin. GND (Pin 4): Ground Pin. PGATE (Pin 5): High Current Gate Driver for External P-Channel MOSFET Switch. Voltage swing is from ground to VIN. VIN (Pin 6): Main Input Voltage Supply Pin. SENSE (Pin 7): Current Sense Input for Monitoring Switch Current. Maximum switch current and Burst Mode threshold is programmed with an external resistor between SENSE and VIN. MODE (Pin 8): Burst Mode Enable/Disable Pin. Connecting this pin to VIN (or above 2V) enables Burst Mode operation, while connecting this pin to ground disables Burst Mode operation. Do not leave floating.
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LTC1771
FUNCTIONAL BLOCK DIAGRA
VIN 1A RUN/SS 1 MODE 8 (BURST ENABLE) 10% CURRENT 1.23V
CSS
+ -
ON EA
VOUT
10% CURRENT
SLEEP
*
ITH 2 RC CC GND 4 1V 2V 1V - + B READY VIN BLANKING
MODE
ON TRIGGER 1-SHOT 3.5s STRETCH VFB 3 VOUT L
*OPTIONAL FOR FOLDBACK
CURRENT LIMITING
4
+
ON
C
-
SOFT-START
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VIN 6 READY 1.23V REFERENCE 22k RSENSE
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VIN CIN
SENSE 7
250k
SW 5
+
COUT
1771 BD
LTC1771
OPERATIO
Main Control Loop The LTC1771 uses a constant off-time, current mode step-down architecture. During normal operation, the P-channel MOSFET is turned on at the beginning of each cycle and turned off when the current comparator C triggers the 1-shot timer. The external MOSFET switch stays off for the 3.5s 1-shot duration and then turns back on again to begin a new cycle. The peak inductor current at which C triggers the 1-shot is controlled by the voltage on Pin 3 (ITH), the output of the error amplifier EA. An external resistive divider connected between VOUT and ground allows EA to receive an output feedback voltage VFB. When the load current increases, it causes a slight decrease in VFB relative to the 1.23V reference, which in turn causes the ITH voltage to increase until the average inductor current matches the new load current. The main control loop is shut down by pulling Pin 1 (RUN/SS) low. Releasing RUN/SS allows an internal 1A current source to charge soft-start capacitor CSS. When CSS reaches 1V, the main control loop is enabled with the ITH voltage clamped at approximately 40% of its maximum value. As CSS continues to charge, ITH is gradually released allowing normal operation to resume. Burst Mode Operation The LTC1771 provides outstanding low current efficiency and ultralow no-load supply current by using Burst Mode operation when the MODE pin is pulled above 2V. During Burst Mode operation, short burst cycles of normal switching are followed by a longer idle period with the switch off and the load current is supplied by the output capacitor. During this idle period, only the minimum required circuitry--1.23V reference and error amp--are left on, and the supply current is reduced to 9A. At no load, the output capacitor is still discharged very slowly by leakage current in the Schottky diode and feedback resistor current resulting in very low frequency burst cycles that add a few more microamps to the supply current.
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(Refer to Functional Block Diagram)
Burst Mode operation is provided by clamping the minimum ITH voltage at 1V which represents about 25% of maximum load current. If the load falls below this level, i.e. the ITH voltage tries to fall below 1V, the burst comparator B switches state signaling the LTC1771 to enter sleep mode. During this time, EA is reduced to 10% of its normal operating current and the external compensation capacitor is disconnected and clamped to 1V so that the EA can drive its output with the lower available current. As the load discharges the output capacitor, the internal ITH voltage increases. When it exceeds 1V the burst comparator exits sleep mode, reconnects the external compensation components to the error amplifier output, and returns EA to full power along with the other necessary circuitry. This scheme (patent pending) allows the EA to be reduced to such a low operating current during sleep mode without adding unacceptable delay to wake up the LTC1771 due to the compensation capacitor on ITH required for stability in normal operation. Burst Mode operation can be disabled by pulling the MODE pin to ground. In this mode of operation, the burst comparator B is disabled and the ITH voltage allowed to go all the way to 0V. The load can now be reduced to about 1% of maximum load before the loop skips cycles to maintain regulation. This mode provides a low noise output spectrum, useful for reducing both audio and RF interference, at the expense of reduced efficiency at light loads. Off-Time The off-time duration is 3.5s when the feedback voltage is close to the reference voltage; however, as the feedback voltage drops, the off-time lengthens and reaches a maximum value of about 70s when VFB is zero. This ensures that the inductor current has enough time to decay when the reverse voltage across the inductor is low such as during short circuit, thus protecting the MOSFET and inductor.
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LTC1771
APPLICATIO S I FOR ATIO
The basic LTC1771 application circuit is shown in Figure 1 on the first page. External component selection is driven by the load requirement and begins with the selection of RSENSE. Once RSENSE is known, L can be chosen. Next, the MOSFET and D1 are selected. The inductor is chosen based largely on the desired amount of ripple current and for Burst Mode operation. Finally CIN is selected for its ability to handle the required RMS input current and COUT is chosen with low enough ESR to meet the output voltage ripple and transient specifications. RSENSE Selection RSENSE is chosen based on the required output current. The LTC1771 current comparator has a maximum threshold of 140mV/RSENSE. The current comparator threshold sets the peak inductor current, yielding a maximum average output current IMAX equal to the peak less half the peak-to-peak ripple current IL. For best performance when Burst Mode operation is enabled, choose IL equal to 35% of peak current. Allowing a margin for variations in the LTC1771 and external components gives the following equation for choosing RSENSE: RSENSE = 100mV/IMAX At higher supply voltages, the peak currents may be slightly higher due to overshoot from current comparator delay and can be predicted from the second term in the following equation:
IPEAK
V -V 0.14 + 0.5 IN OUT RSENSE L(H)
1/ 2
Inductor Value Selection Once RSENSE is known, the inductor value can be determined. The inductance value has a direct effect on ripple current. The ripple current decreases with higher inductance and increases with higher VOUT. The ripple current during continuous mode operation is set by the off-time and inductance to be:
V +V IL(CONT) = t OFF OUT D L
Kool M is a registered trademark of Magnetics, Inc.
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where tOFF = 3.5s. However, the ripple current at low loads during Burst Mode operation is: IL(BURST) 35% of IPEAK 0.05/RSENSE For best efficiency when Burst Mode operation is enabled, choose: IL(CONT) IL(BURST) so that the inductor current is continuous during the burst periods. This sets a minimum inductor value of: LMIN = (70H)(VOUT + VD)(RSENSE) When burst is disabled, ripple currents less than IL(BURST) can be achieved by choosing L > LMIN. Lower ripple current reduces output voltage ripple and core losses, but too low of ripple current will adversely effect efficiency. Inductor Core Selection Once the value of L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool M(R) cores. Actual core loss is independent of core size for a fixed inductor value, but is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates "hard," which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent increase in voltage ripple. Do not allow the core to saturate! Molypermalloy (from Magnetics, Inc.) is a very good, low loss core material for toroids, but it is more expensive than ferrite. A reasonable compromise from the same manufacturer is Kool M. Toroids are space efficient, especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more difficult. However, designs for surface mount are available that do not increase the height significantly.
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LTC1771
APPLICATIO S I FOR ATIO
Power MOSFET Selection
An external P-channel power MOSFET must be selected for use with the LTC1771. The main selection criteria for the power MOSFET are the threshold voltage VGS(TH) and the "on" resistance RDS(ON), reverse transfer capacitance and total gate charge. Since the LTC1771 can operate down to input voltages as low as 2.8V, a sublogic level threshold MOSFET (RDS(ON) guaranteed at VGS = 2.5V) is required for applications that work close to this voltage. When these MOSFETs are used, make sure that the input supply to the LTC1771 is less than the absolute maximum VGS rating (typically 12V), as the MOSFET gate will see the full supply voltage. The required RDS(ON) of the MOSFET is governed by its allowable power dissipation. For applications that may operate the LTC1771 in dropout, i.e. 100% duty cycle, at its worst case the required RDS(ON) is given by: RDS(ON) =
(
PP
IOUT (MAX)
)
2
(1+ P )
where PP is the allowable power dissipation and P is the temperature dependency of RDS(ON). (1 + P) is generally given for a MOSFET in the form of a normalized RDS(ON) vs temperature curve, but = 0.005/C can be used as an approximation for low voltage MOSFETs. In applications where the maximum duty cycle is less than 100% and the LTC1771 is in continuous mode, the RDS(ON) is governed by:
RDS(ON) = DC =
(DC)IOUT2 (1+ P )
VOUT + VD VIN + VD
PP
where DC is the maximum operating duty cycle of the LTC1771. Catch Diode Selection The catch diode carries load current during the off-time. The average diode current is therefore dependent on the P-channel switch duty cycle. At high input voltages the
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diode conducts most of the time. As VIN approaches VOUT the diode conducts only a small fraction of the time. The most stressful condition for the diode is when the output is short-circuited. Under this condition, the diode must safely handle IPEAK at close to 100% duty cycle. To maximize both low and high current efficiencies, a fast switching diode with low forward drop and low reverse leakage should be used. Low reverse leakage current is critical to maximize low current efficiency since the leakage can potentially exceed the magnitude of the LTC1771 supply current. Low forward drop is critical for high current efficiency since loss is proportional to forward drop. The effect of reverse leakage and forward drop on no- load supply current and efficiency for various Schottky diodes is shown in Table 1. As can be seen, these are conflicting parameters and the user must weigh the importance of each spec in choosing the best diode for the application.
Table 1. Effect of Catch Diode on Performance
DIODE MBR0540 UPS5817 MBR0520 MBRS120T3 MBRM120LT3 MBRS320 LEAKAGE NO-LOAD EFFICIENCY (VR = 3.3V) VF @ 1A SUPPLY CURRENT AT 10V/1A 0.25A 2.8A 3.7A 4.4A 8.3A 19.7A 0.50V 0.41V 0.36V 0.43V 0.32V 0.29V 10.4A 11.8A 12.2A 12.2A 14.0A 20.0A 86.3% 88.2% 88.4% 87.9% 89.4% 89.8%
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CIN and COUT Selection At higher load currents, when the inductor current is continuous, the source current of the P-channel MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum capacitor current is given by: CIN required IRMS = IMAX VOUT (VIN - VOUT ) VIN
[
]1/ 2
This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer's
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LTC1771
APPLICATIO S I FOR ATIO
ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Do not underspecify this component. An additional 0.1F ceramic capacitor is also helpful on VIN for high frequency decoupling. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (VOUT) in continuous mode is approximated by:
VOUT
1 IRIPPLE ESR + 8 fCOUT
where f is the operating frequency, COUT is the output capacitance and IRIPPLE is the ripple current in the inductor. For output ripple less than 100mV, assure COUT required ESR is <2RSENSE. The first condition relates to the ripple current into the ESR of the output capacitance while the second term guarantees that the output capacitance does not significantly discharge during the operating frequency period due to ripple current. The choice of using smaller output capacitance increases the ripple voltage due to the discharging term but can be compensated for by using capacitors of very low ESR to maintain the ripple voltage at or below 50mV. The ITH pin OPTI-LOOPTM compensation components can be optimized to provide stable, high performance transient response regardless of the output capacitors selected. Manufacturers such as Nichicon, United Chemicon and Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR for its size of any aluminum electrolytic at a somewhat higher price. Typically once the ESR requirement is satisfied, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. In surface mount applications multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum
OPTI-LOOP is a trademark of Linear Technology Corporation.
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electrolytics and dry tantalum capacitors are both available in surface mount configurations. In case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS, AVX TPSV and KEMET T510 series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. Other capacitor types include Sanyo OS-CON, Sanyo POSCAP, Nichicon PL series and Panasonic SP. Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% - (L1 + L2 +L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in the LTC1771 circuits: the LTC1771 DC bias current, MOSFET gate charge current, I2R losses and catch diode losses. 1. The DC bias current is 9A at no load and increases proportionally with load up to a constant 150A during continuous mode. This bias current is so small that this loss is negligible at loads above a milliamp but at no load accounts for nearly all of the loss. 2. The MOSFET gate charge current results from switching the gate capacitance of the power MOSFET switch. Each time the gate is switched from high to low to high again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is the current out of VIN which is typically much larger than the DC bias current. In continuous mode, IGATECHG = fQP where QP is the gate charge of the internal switch. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 3. I2R losses are predicted from the internal switch, inductor and current sense resistor. In continuous mode the average output current flows through L but is "chopped"
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LTC1771
APPLICATIO S I FOR ATIO
between the P-channel MOSFET in series with RSENSE and the output diode. The MOSFET RDS(ON) plus RSENSE multiplied by the duty cycle can be summed with the resistance of L to obtain I2R losses. 4. The catch diode loss is proportional to the forward drop as the diode conducts current during the off-time and is more pronounced at high supply voltages where the off-time is long. However, as discussed in the Catch Diode section, diodes with lower forward drops often have higher leakage currents, so although efficiency is improved, the no-load supply current will increase. The diode loss is calculated by multiplying the forward voltage drop times the diode duty cycle multiplied by the load current. Other losses including CIN and COUT ESR dissipative losses, and inductor core losses, generally account for less than 2% total additional loss. Output Voltage Programming The output voltage is programmed with an external divider from VOUT to VFB (Pin 1) as shown in Figure 2. The regulated voltage is determined by: VOUT R2 = 1.23 1 + R1
To minimize no-load supply current, resistor values in the megohm range should be used. The increase in supply current due to the feedback resistors can be calculated from:
IVIN = VOUT VOUT R1 + R2 VIN
VOUT
R2 VFB LTC1771 GND
1771 F02
CFF 5pF
R1
Figure 2. LTC1771 Adjustable Configuaration
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A 5pF feedforward capacitor across R2 is recommended to minimize output voltage ripple in Burst Mode operation. Run/Soft-Start Function The RUN/SS pin is a dual purpose pin that provides the soft- start function and a means to shut down the LTC1771. Soft-start reduces the input surge current from VIN by gradually increasing the internal current limit. Power supply sequencing can also be accomplished using this pin. An internal 1A current source charges up an external capacitor CSS. When the voltage on the RUN/SS reaches 1V, the LTC1771 begins operating. As the voltage on the RUN/SS continues to ramp from 1V to 2.2V, the internal current limit is also ramped at a proportional linear rate. The current limits begins near 40% maximum load at VRUN/SS = 1V and ends at maximum load at VRUN/SS = 2.2V. The output current thus ramps up slowly, reducing the starting surge current required from the input power supply. If the RUN/SS has been pulled all the way to ground, there will be a delay before the current limit starts increasing and is given by: tDELAY CSS/ICHG where ICHG 1A. Pulling the RUN/SS pin below 0.5V puts the LTC1771 into a low quiescent current shutdown (IQ < 2A). Foldback Current Limiting As described in the Catch Diode Selection, the worst-case dissipation for diode occurs with a short-circuit output, when the diode conducts the current limit value almost continuously. In most applications this will not cause excessive heating, even for extended fault intervals. However, when heat sinking is at a premium or higher forward voltage drop diodes are being used, foldback current limiting should be added to reduce the current in proportion to the severity of the fault. Foldback current limiting is implemented by adding two diodes in series between the output and the ITH pin as shown in the Functional Diagram. In a hard short (VOUT = 0V) the current will be reduced to approximately 25% of the maximum output current.
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LTC1771
APPLICATIO S I FOR ATIO
Minimum On-Time Considerations
Minimum on-time tON(MIN) is the smallest amount of time that the LTC1771 is capable of turning the top MOSFET on and off again. It is determined by internal timing delays and the amount of gate charge required to turn on the P-channel MOSFET. Low duty cycle applications may approach this minimum on-time limit and care should be taken to ensure that:
V + VD tON = tOFF OUT > tON(MIN) VIN - VOUT
where tOFF = 3.5s and tON(MIN) is generally about 0.5s for the LTC1771. If the duty cycle falls below what can be accommodated by the minimum on-time, the LTC1771 will remain in Burst Mode operation even at high load currents. The output voltage will continue to be regulated, but the ripple current and ripple voltage will increase. Mode Pin Burst Mode operation is disabled by pulling MODE (Pin 8) below 0.5V. Disabling Burst Mode operation provides a low noise output spectrum, useful for reducing both audio and RF interference. It does this by keeping the frequency constant (for fixed VIN) down to much lower load current (1% to 2% of IMAX) and reducing the amount of output voltage and current ripple at light loads. When Burst Mode operation is disabled, efficiency is reduced at light loads and no load supply current increases to 175A. PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1771. These items are also illustrated graphically in the layout diagram of Figure 3. Check the following in your layout:
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1. Is the Schottky diode closely connected to the drain of the external MOSFET and the input cap ground? 2. Is the 0.1F input decoupling capacitor closely connected between VIN (Pin 6) and ground (Pin 4)? This capacitor carries the high frequency peak currents. 3. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1 and R2 must be connected between the (+) plate of COUT and signal ground. Locate the feedback resistors right next to the LTC1771. The VFB line should not be routed close to any nodes with high slew rates. 4. Is the 1000pF decoupling capacitor for the current sense resistor connected as close as possible to Pins 6 and 7? Ensure accurate current sensing with Kelvin connections to the sense resistor. 5. Is the (+) plate of CIN closely connected to the sense resistor ? This capacitor provides the AC current to the MOSFET. 6. Are the signal and power grounds segregated? The signal ground consists of the (-) plate of COUT, Pin 4 of the LTC1771 and the resistive divider. The power ground consists of the Schottky diode anode and the (-) plate of CIN which should have as short lead lengths as possible. 7. Keep the switching node (SW) and the gate node (PGATE) away from sensitive small signal nodes, especially the voltage sensing feedback pin (VFB), and minimize their PC trace area. 8. High impedance nodes such as ITH and VFB are very sensitive to leakage paths on the PC board due to stray flux, solder, epoxy, etc. Make sure PC board is clean. Water-soluble solder flux can be especially leaky if not cleaned properly. Leakage on ITH will manifest itself as excessive output ripple during Burst Mode operation. If the problem persists, adding a 10M resistor from Pin 2 to ground should eliminate the problem.
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LTC1771
APPLICATIO S I FOR ATIO
Design Example
As a design example, assume VIN = 10V (nominal), VIN = 15V(MAX), VOUT = 3.3V, and IMAX = 2A. With this information, we can easily calculate all the important components. RSENSE = 100mV/2A = 0.05 To optimize low current efficiency, MODE pin is tied to VIN to enable Burst Mode operation, thus the minimum inductance necessary is: LMIN = 70H(3.3V + 0.5)(0.05) = 13.3H 15H is chosen for the application.
3.3V + 0.5V IL = 3.5s = 0.89A 15H
For the feedback resistors, choose R1 = 1M to minimize supply current. R2 can then be calculated to be: R2 = (VOUT/1.23 - 1) * R1 = 1.68M Assume that the MOSFET dissipation is to be limited to PP = 0.25W. If TA = 70C and the thermal resistance of the MOSFET is 83C/W, then the junction temperatures will be 91C and P = 0.33. The required RDS(ON) for the MOSFET can now be calculated:
CSS 1 CITH RITH 2 3 4 RUN/SS ITH VFB GND MODE SENSE LTC1771 VIN PGATE 8 7 6 5 MODE
R1
R2
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 3. LTC1771 Layout Diagram
+
CFF
U
P - Channel RDS(ON) = 0.25W 3.3V + 0.5V 2A 10 V + 0.5V = 0.130
W
UU
( ) (1.33)
2
Since the gate of the MOSFET will see the full input voltage, a MOSFET must be selected whose VGS(MAX) > 15V. A P-channel MOSFET that meets both the VGS(MAX) and RDS(ON) requirement is the Si6447DQ. The most stringent requirement for the Schottky diode occurs when VOUT = 0V (i.e., short circuit) at maximum VIN. In this case the worst-case dissipation rises to:
VIN PD = ISC(AVG) (VD) VIN + VD
With a 0.05 sense resistor ISC(AVG) = 2A will result, increasing the 0.5V Schottky diode dissipation to 1W. CIN is chosen for a RMS current rating of at least 1A at temperature. COUT is chosen with an ESR of 0.05 for low output ripple. The output voltage ripple due to ESR is approximately: VORIPPLE (RESR)(IL) = 0.05 (0.89AP-P) = 45mVP-P
+
CIN D1
Q1
0.1F
COUT
L VOUT
1771 F03
11
LTC1771
TYPICAL APPLICATIO S
3.3V to 2.5V/1A Regulator with Burst Mode Operation Enabled
0.01F 1 220pF 10k 2 3 4 RUN/SS ITH VFB GND MODE SENSE LTC1771 VIN PGATE 5 Si3443DV 1M 1% 1.02M 1% 5pF RSENSE 0.1 22H 8 7 6 1000pF VIN 3.3V TO 12V VOUT 2.5V 1A
0.01F 1 220pF 10k 2 3 4 RUN/SS ITH VFB GND MODE SENSE LTC1771 VIN PGATE 5 RSENSE Si6447DQ 0.05 22H 8 7 6 1000pF VIN 5.5V TO 18V VOUT 5V 2A
1M 1%
12
U
+
CIN 22F 25V
+
UPS5817
COUT 150F 6.3V
1771 TA01
5V/2A Regulator with Burst Mode Operation Disabled
+
3.09M 1% 5pF
CIN 22F 25V
+
UPS5817
COUT 150F 6.3V
1771 TA04
LTC1771
TYPICAL APPLICATIO S
0.01F 1 220pF 10k 2 3 4 RUN/SS ITH VFB GND MODE SENSE LTC1771 VIN PGATE 5 RSENSE 0.05 15H 8 7 6 1000pF MODE
1M 1%
0.01F 1 220pF 10k 2 3 4 1M 1% RUN/SS ITH VFB GND MODE SENSE LTC1771 VIN PGATE 8.66M 1% 5pF 5 Si6435DQ RSENSE 0.025 8 7 6 1000pF VIN 5V TO 18V MODE
U
Low Dropout Single Cell Lithium-Ion to 3V
+
Si3443DV 1.43M 1% 5pF UPS5817
CIN 22F 25V
Li-Ion 3.3V TO 4.2V VOUT 3V 2A
+
COUT 150F 6.3V
1771 TA02
12V/1A Zeta Converter
+
CIN 22F 25V
*
47H
*
47H
22F 20V
+
UPS5817
COUT 150F 20V
1771 TA05
VOUT 12V 1A
+
13
LTC1771
TYPICAL APPLICATIONS
0.01F 1 220pF 10k 2 3 4 RUN/SS ITH VFB GND MODE SENSE LTC1771 VIN PGATE 1.02M 1% 5pF U1 56 7 8 22H 1N4148 x2 VOUT 2.5V 1A 5 RSENSE 0.1 43 2 8 7 6 1000pF VIN 2.8V TO 12V MODE
1M 1%
U1: INTERNATIONAL RECTIFIER FETKY TM IRF7422D2
0.01F 1 220pF 10k 2 3 4 RUN/SS ITH VFB GND MODE SENSE LTC1771 VIN PGATE 5 Si6447DQ 1M 1% 4.69M 1% 5pF RSENSE 0.1 47H 8 7 6 1000pF VIN 8V CIN TO 18V 22F 25V UPS5817 VOUT 4-NiCd 1A COUT 100F 10V
1771 TA07
14
U
2.5V/1A Regulator with Foldback Current Limit
+
1
CIN 22F 25V ITH
+
COUT 150F 6.3V
1771 TA06
4-NiCd Battery Charger
MODE
+
+
UPS5817
LTC1771
PACKAGE DESCRIPTIO
0.007 (0.18) 0.021 0.006 (0.53 0.015)
* DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE ** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
0.010 - 0.020 x 45 (0.254 - 0.508) 0.008 - 0.010 (0.203 - 0.254) 0- 8 TYP
0.014 - 0.019 (0.355 - 0.483) TYP *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
0.016 - 0.050 (0.406 - 1.270)
U
Dimension in inches (millimeters) unless otherwise noted. MS8 Package 8-Lead Plastic MSOP
(LTC DWG # 05-08-1660)
0.118 0.004* (3.00 0.102)
8
76
5
0.193 0.006 (4.90 0.15)
0.118 0.004** (3.00 0.102)
1 0.040 0.006 (1.02 0.15) 0 - 6 TYP SEATING PLANE 0.012 (0.30) 0.0256 REF (0.65) BSC
23
4 0.034 0.004 (0.86 0.102)
0.006 0.004 (0.15 0.102)
MSOP (MS8) 1098
S8 Package 8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 - 0.197* (4.801 - 5.004) 8 7 6 5
0.228 - 0.244 (5.791 - 6.197)
0.150 - 0.157** (3.810 - 3.988)
1
2
3
4
0.053 - 0.069 (1.346 - 1.752)
0.004 - 0.010 (0.101 - 0.254)
0.050 (1.270) BSC
SO8 1298
15
LTC1771
TYPICAL APPLICATIO
0.01F 1 220pF 10k 2 3 4 1M 1% RUN/SS ITH VFB GND MODE SENSE LTC1771 VIN PGATE 3.09M 1% 5pF 5 Si3443DV 8 7 6 1000pF MODE
RELATED PARTS
PART NUMBER LTC1147 Series LTC1174/LTC1174-3.3/LTC1174-5 LTC1265 LTC1474/LTC1475 LTC1574/LTC1574-3.3/LTC1574-5 LTC1622 LTC1624 LT(R)1761 Series LT1763 Series LTC1772 DESCRIPTION High Efficiency Step-Down Switching Regulator Controllers High Efficiency Step-Down and Inverting DC/DC Converters 1.2A High Efficiency Step-Down DC/DC Converter Low Quiscent Current Step-Down Regulators High Efficiency Step-Down DC/DC Converters with Internal Schottky Diode Low Input Voltage Step-Down DC/DC Controller High Efficiency SO-8 N-Channel Switching Regulator Controller 100mA, Low Noise, LDO Micropower Regulators in SOT-23 500mA, Low Noise, LDO Micropower Regulators Constant Frequency Step-Down DC/DC Controller COMMENTS 100% DC, 3.5V VIN 16V Selectable IPEAK = 300mA or 600mA Burst Mode Operation, Internal MOSFET Monolithic, IQ = 10A, 400mA, MS8 LTC1174 with Internal Schottky Diode Constant Frequency, 2V to 10V VIN, MS8 95% DC, 3.5V to 36V VIN 20A Quiescent Current, 20VRMS Noise 30A Quiescent Current, 20VRMS Noise SOT-23, 2.2V to 9.8V VIN
16
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 q FAX: (408) 434-0507 q www.linear-tech.com
U
5V/1A Zeta Converter
VIN (V) ILOAD(MAX) (A) 2.8 0.8 3.3 1.1 5 1.7 7.5 2.3 10 2.7 12 2.9 VIN 2.8V TO 12V RSENSE 0.025
+
CIN 22F 25V
*
22H
*
22H
22F 10V
+
UPS5817
COUT 150F 6.3V
1771 TA03
VOUT 5V 1A
+
1771i LT/TP 0200 4K * PRINTED IN USA
(c) LINEAR TECHNOLOGY CORPORATION 2000

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